1. Field of the Invention
This invention relates to multi-layer ceramic lowpass filters particularly for use in mobile communication instruments such as portable telephones and cordless telephones.
2. Description of Related Art
The following references provide background information relating to multi-layer ceramic lowpass and are hereby incorporated by reference in their entireties:
[1] D. Swanson, "Thin-Film Lumped-Element Microwave Filters," 1989 IEEE MTT-S Digest, pages 671-674, 1989. PA1 [2] J. Helszajn, "Microwave Planar Passive Circuits and Filters," Chapter 15, John Wiley & Sons, 1994. PA1 [3] M. Miyazaky, et al., "A Broad Band Dielectric Diplexer Using a Snake Strip-Line," 1991 IEEE MTT-S Digest, pages 551-554, 1991. PA1 [4] U.S. Pat. No. 5,357,227 to Tinegawa, et al., 1994.
Depending on their implementation, filters can be grouped into three types: lumped-type, distributed-type and semi-lumped-type (which are constructed by lumped elements and distributed elements). The size of the distributed element, which is usually a section of transmission line, is determined by the signal wavelength associated with the operational frequencies. As a result, at frequencies lower than 200 MHz, the implementation of a filter by semi-lumped type or distributed-type elements is impractical due to the unacceptably large size of the distributed elements. However, in the higher microwave frequency region or millimeter wave frequency region, the distributed-type filters are always used due to acceptable size and better performance than the lumped-type filters. As discussed in reference [1], the reasons why the lumped elements are not suitable to be used in filter designs at frequencies higher than several hundred MHz are that the lumped inductors are too lossy and the parasitic effects make the control of filter performance more difficult. Although the problem of losses in lumped elements can be improved in superconductor applications, such applications are limited.
At frequencies between several hundred MHz to several Ghz, the size of the distributed elements is not appreciated in mobile communication instruments. The trends in developing mobile communication instruments are miniaturization and power saving. There are two ways to design a miniaturized filter with high performance; one is to employ a semi-lumped configuration and the other is to use a high dielectric constant structure. The semi-lumped type filters usually employ chip capacitors, inter-digital type capacitors or metal-insulation-metal (MIM) capacitors, which are not as lossy as lumped inductors, and some sections of distributed transmission lines, which are usually much shorter than 1/4 signal wavelength. In addition to the capacity with regard to miniaturization of the semi-lumped configuration, this type of filter also has the ability to suppress periodic spurious signals from which the distributed type filters usually suffer. Recently, the high dielectric constant ceramic filters, such as coaxial type or mono-block type, are very popular in the frequency region of several hundred MHz to several Ghz due to high performance and small size. The high performance results from the shape of the cross-section of the transmission line which is usually round or smoothly curved, yielding lower conductor losses. The small size is enabled because the dielectric constant of ceramic materials is very high, resulting in a reduction in the signal wavelength. Yet, such kinds of filters are almost always used with regard to applications of bandpass filters or bandstop filters.
FIG. 1 is a perspective view of a conventional high frequency lowpass filter as described in reference [2]. In FIG. 1, the narrow microstrip transmission line sections 4a, 4b are used as equivalent series inductors. The wide transmission line sections 5a, 5b, 5c are used as equivalent capacitors connected to ground plane 3. More particularly, a first capacitive open-circuited stub electrode 5a forming a part of a first capacitor and an input electrode 6a forming an input terminal extend from the end of microstrip line electrode 4a. A second capacitive open-circuited stub electrode 5b forming part of a second capacitor extends from the other end of microstrip line electrode 4a and one end of the other end of the other microstrip line electrode 4b. A third capacitive open-circuited stub electrode 5c forming part of a third capacitor and an output electrode 6b as an output terminal extend from the other microstrip line electrode 4b. The equivalent circuit is as shown in FIG. 2.
The drawbacks of this filter are as follows. The filter order is high in general applications resulting from the attenuation poles of the filter all being located at infinite frequency. This in turn results in a larger circuit size. The filter order is defined from the numbers of the branches in the equivalent circuit. One branch is defined as an inductor, a capacitor, a series-connected capacitor and inductor or a shunt-connected capacitor and inductor. Since the branches in FIG. 2 are only single-capacitor or single-inductor, these branches contribute attenuation poles at infinite frequency. The attenuation pole means that all of the signal can not pass the filter. A series inductor is equivalent to an open circuit to infinite frequency (j.omega.L.fwdarw..infin., if .omega..fwdarw..infin.). A capacitor connected to ground is equivalent to a short circuit at infinite frequency (1/j.omega.C.fwdarw.0, if .omega..fwdarw..infin.). When a signal meets an open circuit or a short circuit, the total signal will be reflected back to the feed and no pass occurs. The number of attenuation poles at infinite frequency determine the slope of the rejection curve at the stopband. The larger the number is, the sharper the slope will be. Additionally, if the filter in FIG. 1 is used in the RF band, the capacitors in FIG. 2 usually have large values. This results in the areas occupied by the wide line sections 5a, 5b, 5c in FIG. 1 being necessarily large, since in general the substrate thickness can not be too thin due to requirements for the supporting strength of the total circuit. Therefore, this kind of filter is too large in size for low frequency applications. Also, if the lengths of the wide line sections 5a, 5b, 5c in FIG. 1 are decreased with the capacitor values being kept unchanged, then the widths of the wide line sections 5a, 5b, 5c must become even greater. This tradeoff means that it is quite difficult for this type of filter to be miniaturized. Further, because the size of the wide line sections 5a, 5b, 5c in FIG. 1 is too large, these pads (the wide line sections 5a, 5b, 5c) are not much smaller than the wavelength, and the resonance phenomena will occur in the higher frequency range. This results in spurious response in the higher frequency range and degradation of the stopband performance.
FIG. 3 shows another example from reference [2]. The equivalent circuit of this filter is shown in FIG. 4. In FIG. 3, narrow microstrip lines 34a, 34b are used as equivalent series inductors. Wide transmission line sections 35a, 35b, 35c are used as equivalent capacitors connected to ground plane 3. Narrow microstrip lines 34c, 34d, 34e are used as equivalent inductors in series with equivalent capacitors 35a, 35b, 35c, respectively. Input electrode 36a functions as an input terminal, and output electrode 36b functions as an output terminal. Comparing the equivalent circuit of FIG. 2 with that of FIG. 4, the only difference is the capacitor branch is changed into a series-connected capacitor and inductor. The branch formed of the series-connected capacitor and inductor elements can contribute an attenuation pole at the resonant frequency of these elements. This branch is equivalent to a short circuit to ground at resonant frequency. This indicates that this type of filter can improve the high filter order problem incurred by the FIG. 2 device. However, although the FIG. 3 filter can improve the high order problem, the element size is still too large. Therefore, the higher frequency performance is still poor.
FIG. 5 shows a snake-type lowpass filter described in reference [3]. Its equivalent circuit is shown in FIG. 6. In FIG. 5, narrow microstrip lines 54a-54g are used as equivalent inductors. Wide transmission line sections 55a-55d are located side-by-side and are used as equivalent capacitors connected to ground plane 3. The side-by-side equivalent capacitors 55a-55d produce capacitances between adjacent equivalent capacitors which are in parallel with equivalent inductors 54b-54f. Input electrode 56a functions as an input terminal, and output electrode 56b functions as an output terminal. In this circuit, there are some branches formed of shunt-connected capacitor and inductor pairs. These branches also contribute attenuation poles at their resonant frequencies and are equivalent to an open circuit at their resonant frequencies. Although this type of filter can have attenuation poles at a finite frequency, the attenuation poles cannot be located at the neighboring region of the passband since the capacitor in the shunt capacitor and inductor pair is too small (they are side-coupled as shown in FIG. 5). Therefore, in the FIG. 5 device, the capacity to reduce the filter order is limited.